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ABSTRACT
DC Motor speed control is carried out by use of Four Quadrant Chopper drive. Insulated Gate
Bipolar Transistors are used for speed control of the motor and the IGBT triggering is carried out
by use of PWM converters under various loading conditions and by varying armature voltage
and field voltage. The above mentioned experiment was again carried out using Thyristors and a
comparative study was made.
INTRODUCTION
DC motors are used extensively in adjustable-speed drives and position control applications.
Their speeds below the base speed can be controlled by armature-voltage control. Speeds above
the base speed are obtained by field-flux control. As speed control method for DC motors are
simpler and less expensive than those for the AC motors, DC motors are preferred where widespeed
range control is required. DC choppers also provide variable dc output voltage from a
fixed dc input voltage.
The Chopper circuit used can operate in all the four quadrants of the V-I plane. The output
voltage and current can be controlled both in magnitude as well as in direction so the power flow
can be in either direction. The four-quadrant chopper is widely used in reversible dc motor
drives. By applying chopper it is possible to implement regeneration and dynamic braking for dc
motors
EXPERIMENTAL DETAILS
The experimental set up consists of a Four-quadrant IGBT based chopper driver model
PEC-16HV3 .The ac supply in fed to the setup through an isolation transformer and it is rectified
to dc for its use. The PWM converters generate pulse-modulated signal that are compared with
the base signal and are fed to OPTO. Delay logic is provided to gate drivers and thus the signal
obtained is the gating signal for the IGBTs in the four-quadrant chopper. Once the IGBTs are
triggered they are used in pairs to control the speed of the dc motor.
DC motor Specification: 220V, 2.2A, 1420 rpm Shunt type single phase.
Another set up consisted of half bridge rectifier consisting of thyristor wherein the speed control
for the same DC motor was carried out using the firing angle of the thyristor.
CONCLUSION
1. Speed varies directly with armature voltage by keeping field voltage constant.
2. Speed varies inversely with field voltage by keeping armature voltage constant.
3. Armature voltage control gives the speed below the base speed whereas field control
gives the speed control above the base speed.
4. Armature current vs. Speed at constant flux gives a drooping characteristic. Though it
should have been a straight line parallel to x-axis but due to saturation effect there is
slight decrease in speed and shows a drooping characteristic.
5. The IGBT based circuit gives smoother control over the entire speed range as compared
with the SCR based circuit.
The above conclusions were found to be in accordance with the theoretical results.
REFERENCES
1. Rashid, Muhammad H. Power Electronics. New Delhi: Prentice Hall of India Pvt Ltd,
2001.
2. Bimbhra, Dr P S. Power Electronics. New Delhi: Khanna Publisher, 2005.
3. Bimbhra, Dr P S. Electrical Machinery. New Delhi: Khanna Publisher, 1998.
4. Pendharkar Sameer, Trivedi Malay, Shenai Krishna,” Electrothermal Simulations in
Punchthrough and Nonpunch through IGBT’s”, IEEE transactions on electron devices,
Vol. 45, no. 10, october 1998.
5. Yilmaz H., VanDell W R., Owyang K, and Chang M. F, “Insulated gate transistor
modeling and optimization,” in IEDM Tech. Dig., 1984, p. 274.
INTRODUCTION
The chopper circuit shown in fig.1 can operate in all four quadrants of the Vo-Io plane. That is
the output voltage and current can be controlled both in magnitude and direction. Therefore, the
power flow can be in any direction.
In the first quadrant the power flows from the source to the load and is assumed to be
(+ve).
In the second quadrant, the voltage is still positive but the current is negative. Therefore,
the power is negative. In this case, the power flows from load to source and this can happen if
the load is inductive or back emf source such as a dc motor.
In the third quadrant both the voltage and current are negative but the power is positive.
In the fourth quadrant voltage is negative but current is positive. The power is therefore
negative.
This chopper is widely used in reversible dc motors drives. The reversible dc motor
drive requires power flow in either direction in order to achieve fast dynamics response. By
employing four-quadrant chopper it is possible to implement regeneration and dynamic braking
by means of which fast dynamic response is achieved.
CIRCUIT DESCRIPTION
The four quadrant chopper with four switching devices where diodes are connected in anti
parallel with the switching devices is also referred to as full bridge converter topology. The input
to the full bridge converter is fixed magnitude dc voltage Vdc. The output of the converter can be
a variable dc voltage with either polarity. The circuit is therefore called as four quadrant chopper
circuit or dc to dc converter. The output of the full bridge converter can also be an ac voltage
with variable frequency and amplitude in which case the converter is called as dc- to-ac
conversion ( inverter). In a full bridge converter when a gating signal is given to a switching
device either the switching device or the diode only will conduct depending on the directions of
the output load current.
SWITCHING MODES OF FOUR QUADRANT CHOPPER
The switches in the four quadrant chopper can be switched in two different modes such that:
• The output voltage swings in both direction i.e. from +Vdc to –Vdc. This mode of
switching is referred to as PWM with bipolar voltage switching.
• The output voltage swings either from –zero to +Vdc or zero to- Vdc. This mode of
switching is referred to as PWM with unipolar voltage switching.
OPERATION OF THE FOUR QUADRANT CHOPPER WITH BIPOLAR VOLTAGE
SWITCHING
The operation of the circuit as a four quadrant chopper with bipolar voltage switching is
explained, referring to the circuit diagram of Fig 1.2. When the switches T1 and T4 are turned
ON by applying gating signals simultaneously, the load voltages Vdc with terminal ‘A’ positive
and the load current IL flows in the direction from A to B. Because of the load inductance, the
current cannot change instantaneously.
The load voltage V will now be – Vdc since the conduction of the diode D3 will connect the load
terminal B to the (+) ve terminal of the source. As the load voltage is negative and the current is
still positive, the power is negative. The power now flows from the load to the source. This
corresponds to the operation of chopper circuit in the fourth quadrant. This operation in the fouth
quadrant will continue as long as the current is positive. When T1 and T4 are off, T3 and T2 can
be turned ON.
When the current passes through zero, the devices T3 and T2 can be turned on, and
the load current becomes negative. The load current now passes through T3 and T2 with current
direction in the load as from B to A. this brings the operation of the chopper in the third
quadrant. Turning of the T3 and T2 will bring in the conduction of the diode D1 and D4 and the
operation of the chopper circuit in the second quadrant.
The operation of the chopper in the first and third quadrant corresponds to power flow from the
source to the load, and is considered to be forward power flow. The operation in the fourth and
second quadrant corresponds to reverse power flow. The relevant waveforms showing the
operation of full bridge converter in all the four quadrant
GENERATION OF GATING SIGNALS
The gating signals for the switches in the four quadrant chopper are derived by comparing a
triangular wave with a control voltage level. The generation of gating signals for a unipolar
voltage switching is shown in fig 1.4.
The triangular carrier waveform is compared with the control voltage (+)v and (-)v. the pulse
generated by comparing +v with triangular carrier is used to turn on T1and its compliment is
used to turn on T2. The pulse generated by comparing -ve with triangular carrier is used to turn
on T3 and its complement is used to turn on T4. The voltage varies from –Vtri to +Vtri.
The fig. below shows the schematic of the generator of gating signal for the four quadrant
chopper with unipolar switching. A triangular carrier wave of frequency around 2 Khz is generated . The triangular wave is compared with +Vc and –Vc in comparator 1 and comparator 2 respectively.
HARDWARE DESCRIPTION
The hardware involved in the four quadrant chopper drive is screen printed on the front panel .it
consists of both the power circuitry and the control circuitry.
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POWER CIRCUIT
It consists ofi)
single phase diode bridge rectifier
ii) four quadrant chopper
iii) DC link capacitors
iv) Braking circuit
v) Field control chopper
vi) EMI filter.
The diode rectifier rectifies the input ac voltage and provides the dc voltage to the chopper.
Large values of dc link capacities maintain a constant dc voltage is also used for the field circuit
of the motor through a single quadrant chopper.
The chopper consists of four IGBTs rated at 900V, 60A.
IGBT BASICS
2.1 INTRODUCTION
Recent technology advances in power electronics have arisen primarily from improvements in
semiconductor power devices, with insulated gate bipolar transistors (IGBT) leading the market
today for medium power applications. IGBTs feature many desirable properties including a MOS
input gate, high switching speed, low conduction voltage drop, high current carrying capability,
and a high degree of robustness. Devices have drawn closer to the 'ideal switch', with typical
voltage ratings of 600 - 1700 volts, on-state voltage of 1.7 - 2.0 volts at currents of up to 1000
amperes, and switching speeds of 200 - 500 ns. The availability of IGBTs has lowered the cost of
systems and enhanced the number of economically viable applications. The insulated gate
bipolar transistor (IGBT) combines the positive attributes of BJTs and MOSFETs. BJTs have
lower conduction losses in the on-state, especially in devices with larger blocking voltages, but
have longer switching times, especially at turn-off while MOSFETs can be turned on and off
much faster, but their on-state conduction losses are larger, especially in devices rated for higher
blocking voltages. Hence, IGBTs have lower on-state voltage drop with high blocking voltage
capabilities in addition to fast switching speeds.
IGBTs have a vertical structure as shown in Fig. 2.1. This structure is quite similar to that of the
vertical diffused MOSFET except for the presence of the p+ layer that forms the drain of the
IGBT. This layer forms a p-n junction (labeled J1 in the figure), which injects minority carriers
into what would appear to be the drain drift region of the vertical MOSFET. The gate and source
of the IGBT are laid out in an inter-digitated geometry similar to that used for the vertical MOSFET.
TURN-ON TRANSIENTS
The turn-on switching transient of an IGBT with an inductive load is shown in Fig. 2.4. The
turn-on switching transients of IGBTs are very similar to MOSFETs since the IGBT is
essentially acting as a MOSFET during most of the turn-on interval. With gate voltage applied
across the gate to emitter terminals of the IGBT, the gate to emitter voltage rises up in an
exponential fashion from zero to VGE(th) due to the circuit gate resistance (RG) and the gate to
emitter capacitance (Cge). The Miller effect capacitance (Cgc) effect is very small due to the high
voltage across the device terminals.
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Beyond VGE(th), the gate to emitter voltage continues to rise as before and the drain current begins
to increase linearly as shown above. Due to the clamp diode, the collector to emitter voltage
remains at Vdc as the IGBT current is less than Io. Once the IGBT is carrying the full load
current but is still in the active region, the gate to emitter voltage becomes temporarily clamped
to VGE,Io, which is the voltage required to maintain the IGBT current at Io. At this stage, the
collector to emitter voltage starts decreasing in two distinctive intervals tfv1 and tfv2. The first
time interval corresponds to the traverse through the active region while the second time interval
corresponds to the completion of the transient in the ohmic region.
During these intervals, the Miller capacitance becomes significant where it discharges to
maintain the gate to source voltage constant. When the Miller capacitance is fully discharged, the
gate to emitter voltage is allowed to charge up to VG and the IGBT goes into deep saturation. The
resultant turn on switching losses are shown in the above figure. The on energy loss is
approximately estimated via,
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The above switching waveforms are ideal in the since that the clamp diode reverse recovery
effects are neglected. If these effects are included, an additional spike in the current waveform
results as shown in the previous figure. As a result, additional energy losses will be incurred
within the device.
2.4 TURN-OFF TRANSIENTS
The turn-off switching transients of an IGBT with an inductive load are shown in Fig. 2.5. When
a negative gate signal is applied across the gate to emitter junction, the gate to emitter voltage
starts decreasing in a linear fashion. Once the gate to emitter voltage drops below the threshold
voltage (VGE(th)), the collector to emitter voltage starts increasing linearly. The IGBT current
remains constant during this mode since the clamp diode is off. When the collector to emitter
voltage reaches the dc input voltage, the clamp diode starts conducting and the IGBT current
falls down linearly. The rapid drop in the IGBT current occurs during the time interval tfi1, which
corresponds, to the turn-off of the MOSFET part of the IGBT (Fig. 2.5). The tailing of the
collector current during the second interval tfi2 is due to the stored charge in the n- drift region of
the device. This is because the MOSFET is off and there is no reverse voltage applied to the
IGBT terminals that could generate a negative drain current so as to remove the stored charge.
The only way for stored charge removal is by recombination within the n- drift region. Since it is
desirable that the excess carriers lifetime be large to reduce the on-state voltage drop, the
duration of the tail current becomes long. This will result in additional switching losses within
the device. This time increases also with temperature similar to the tailing effect in BJTs. Hence,
a trade off between the on-state voltage drop and faster turn-off times must be made.